LED drive circuit with a programmable input for LED lighting

ABSTRACT

A LED drive circuit according to the present invention comprises a controller and a programmable signal. The controller generates a switching signal coupled to switch a magnetic device for generating an output current to drive a plurality of LEDs. The programmable signal is coupled to regulate a current-control signal of the controller. The switching signal is modulated in response to the current-control signal for regulating the output current, and the level of the output current is correlated to the current-control signal.

REFERENCE TO RELATED APPLICATIONS

This Application is based on Provisional Patent Application Ser. No. 61/335,749, filed 11 Jan. 2010, currently pending.

BACKGROUND OF THE INVENTION

1. Field of Invention

The present invention relates to a LED lighting, and more particularly, the present invention relates to a switching regulator with programmable input.

2. Description of Related Art

The LED driver is used to control the brightness of the LED in accordance with its characteristics. The LED driver is also utilized to control the current that flows through the LED. The present invention provides a primary-side controlled switching regulator with a programmable input for a LED driver. One object of this invention is to improve the power factor (PF) of the LED driver. The programmable input can also be used for the dimming control. It is another object of the invention.

SUMMARY OF THE INVENTION

It is an objective of the present invention to provide a LED drive circuit with programmable input. It can modulate the switching signal to regulate the output current for improving the power factor (PF) of the LED drive circuit.

It is an objective of the present invention to provide a LED drive circuit with programmable input. The programmable input can be used for the dimming control.

The LED drive circuit according to the present invention comprises a controller and a programmable signal. The controller generates a switching signal coupled to switch a magnetic device for generating an output current to drive a plurality of LEDs. The programmable signal is coupled to regulate a current-control signal of the controller. The switching signal is modulated in response to the current-control signal for regulating the output current. The level of the output current is correlated to the current-control signal. Further, the programmable signal is coupled to control a reference signal of a current-loop of the controller. The switching signal is modulated in response to the reference signal. The level of the output current is correlated to the reference signal.

The LED driver according to the present invention comprises a controller and a programmable signal. The controller generates a switching signal coupled to switch a transformer for generating a current input signal coupled to the controller and an output current connected to drive a plurality of LEDs. The programmable signal is coupled to modulate the current input signal. The current input signal is further coupled to generate a current-control signal. The current input signal is correlated to a switching current of the transformer. The switching signal is controlled in response to the current-control signal. The level of the output current is correlated to the current-control signal.

BRIEF DESCRIPTION OF ACCOMPANIED DRAWINGS

The accompanying drawings are included to provide a further understanding of the present invention, and are incorporated in and constitute a part of this specification. The drawings illustrate embodiments of the present invention and, together with the description, serve to explain the principles of the present invention. In the drawings,

FIG. 1 shows a circuit diagram of a preferred embodiment of a LED drive circuit in accordance with the present invention.

FIG. 2 is another preferred embodiment of the LED drive circuit in accordance with the present invention.

FIG. 3 is a preferred embodiment of the controller in accordance with the present invention.

FIG. 4 is a preferred embodiment of the integrator in accordance with the present invention.

FIG. 5 shows a preferred embodiment of the maximum duty circuit in accordance with the present invention.

FIG. 6 is another preferred embodiment of the controller in accordance with the present invention.

FIG. 7 shows a preferred embodiment of the voltage-to-current converter in accordance with the present invention.

DETAILED DESCRIPTION OF PREFERRED EMBODIMENTS

FIG. 1 is a circuit diagram of a preferred embodiment of a LED drive circuit in accordance with the present invention. The LED drive circuit, which is presently preferred to be a LED circuit or a LED driver. An offline transformer 10 is a magnetic device including a primary winding N_(P), an auxiliary winding N_(A) and a secondary winding N_(S). One terminal of the primary winding N_(P) is coupled to receive an input voltage V_(IN). The other terminal of the primary winding N_(P) is coupled to a drain terminal of a power transistor 20. The power transistor 20 is utilized to switch the offline transformer 10. One terminal of the secondary winding N_(S) connects one terminal of a rectifier 40. A filter capacitor 45 is coupled between the other terminal of the rectifier 40 and the other terminal of the secondary winding N_(S). A plurality of LEDs 101 . . . 109 are connected in series and connected to the filter capacitor 45 in parallel.

A controller 70 comprises a supply terminal VCC, a voltage-detection terminal VDET, a ground terminal GND, a current-sense terminal VS, an input terminal VCNT and an output terminal VPWM. The controller 70 is a primary-side controller that is coupled to control the power transistor 20 for switching the primary winding N_(P) of the magnetic device. The voltage-detection terminal VDET is coupled to the auxiliary winding N_(A) via a resistor 50 to receive a voltage-detection signal V_(DET) for detecting a reflected voltage V_(AUX). The voltage-detection signal V_(DET) is correlated to the reflected voltage V_(AUX). The reflected voltage V_(AUX) further charges a capacitor 65 via a rectifier 60 for powering the controller 70. The capacitor 65 is coupled to the supply terminal VCC of the controller 70.

The current-sense terminal VS is coupled to a current-sense resistor 30. The current-sense resistor 30 is coupled from a source terminal of the power transistor 20 to a ground for converting a switching current I_(P) of the magnetic device to a current input signal V_(IP). The switching current I_(P) flows the power transistor 20. The output terminal VPWM outputs a switching signal V_(PWM) to switch the offline transformer 10. The controller 70 generates the switching signal V_(PWM) to switch the magnetic device through the power transistor 20 for generating an output current I_(O) and controlling the switching current I_(P). The output current I_(O) is coupled to drive LEDs 101 . . . 109. The input terminal VCNT receives a programmable signal V_(CNT) to control the switching current I_(P) and the output current I_(O).

FIG. 2 is another preferred embodiment of the LED drive circuit in accordance with the present invention. Comparing with FIG. 1 and FIG. 2, the primary winding N_(P) is coupled to receive the input voltage V_(IN) rectified and filtered by a bridge rectifier 80 and a bulk capacitor 89 from an AC input V_(AC). The AC input V_(AC) is coupled to an input of the bridge rectifier 80. The bulk capacitor 89 is coupled between an output of the bridge rectifier 80 and the ground. Moreover, the programmable signal V_(CNT) is generated at the input terminal VCNT in response to the AC input V_(AC) of the LED drive circuit through diodes 81 and 82, a voltage divider formed by resistors 85 and 86, a filter capacitor 87. Anodes of the diodes 81 and 82 are coupled to receive the AC input V_(AC). One terminal of the resistor 85 is coupled to cathodes of the diodes 81 and 82. The resistor 86 is connected between the other terminal of the resistor 85 and the ground. The filter capacitor 87 is connected to the resistor 86 in parallel. The filter capacitor 87 is further coupled to the input terminal VCNT. Other circuits of this embodiment are the same as the embodiment of FIG. 1, so here is no need to describe again.

FIG. 3 is a preferred embodiment of the controller in accordance with the present invention. The controller 70 is a primary-side controller coupled to switch the primary winding N_(P) of the offline transformer 10. The detail description of the primary-side controlled regulator can be found in a prior art “Control circuit for controlling output current at the primary side of a power converter” U.S. Pat. No. 6,977,824.

A waveform detector 300 detects the witching current I_(P) (as shown in FIG. 1) and generates current-waveform signals V_(A) and V_(B) by sampling the current input signal V_(IP) through the current-sense terminal VS. The waveform detector 300 further receives the switching signal V_(PWM), a pulse signal PLS and a clear signal CLR. A discharge-time detector 100 receives the voltage-detection signal V_(DET) via the auxiliary winding N_(A) (as shown in FIG. 1) to detect the discharge-time of a secondary side switching current I_(S) and generate a discharge-time signal S_(DS). The secondary side switching current I_(S) is proportional to the switching current I_(P). The pulse width of the discharge-time signal S_(DS) is correlated to the discharge-time of the secondary side switching current I_(S). The output current I_(O) is correlated to the secondary side switching current I_(S). An oscillator (OSC) 200 generates the pulse signal PLS coupled to a PWM circuit 400 to determine the switching frequency of the switching signal V_(PWM). The oscillator 200 further generates the clear signal CLR that is coupled to the waveform detector 300 and an integrator 500.

The integrator 500 is used to generate a current signal V_(Y) by integrating an average current signal I_(AVG) (as shown in FIG. 4) with the discharge-time signal S_(DS). The average current signal I_(AVG) is produced in response to the current-waveform signals V_(A) and V_(B). A time constant of the integrator 500 is correlated with a switching period T of the switching signal V_(PWM). The current signal V_(Y) is therefore related to the output current I_(O). An operational amplifier 71 and a reference signal V_(REF1) develop an error amplifier for output current control. A positive input of the operational amplifier 71 is coupled to receive the reference signal V_(REF1). A negative input of the operational amplifier 71 is coupled to receive the current signal V_(Y). The error amplifier amplifies the current signal V_(Y) and provides a loop gain for output current control.

A comparator 75 is associated with the PWM circuit 400 for controlling the pulse width of the switching signal V_(PWM) in response to an output of the error amplifier. A positive input and a negative input of the comparator 75 are coupled to receive the output of the error amplifier and a ramp signal RMP respectively. The ramp signal RMP is provided by the oscillator 200. An output of the comparator 75 generates a current-control signal S_(I) for controlling the pulse width of the switching signal V_(PWM). A current control loop is formed from detecting the switching current I_(P) to modulate the pulse width of the switching signal V_(PWM). The current control loop controls the magnitude of the switching current I_(P) in response to the reference signal V_(REF1).

The PWM circuit 400 outputs the switching signal V_(PWM) for switching the offline transformer 10. The PWM circuit 400 according to one embodiment of the present invention comprises a D flip-flop 95, an inverter 93, an AND gate 91 and an AND gate 92. A D input of the D flip-flop 95 is supplied with a supply voltage V_(CC). An output of the inverter 93 is coupled to a clock input CK of the D flip-flop 95. The pulse signal PLS sets the D flip-flop 95 through the inverter 93. An output Q of the D flip-flop 95 is coupled to a first input of the AND gate 92. A second input of the AND gate 92 is coupled to the output of the inverter 93 and receives the pulse signal PLS through the inverter 93. An output of the AND gate 92 is also an output of the PWM circuit 400 that generates the switching signal V_(PWM). The D flip-flop 95 is reset by an output of the AND gate 91.

A first input of the AND gate 91 is supplied with a voltage-control signal S_(V). The voltage-control signal S_(V) is generated by a voltage control loop, in which the voltage control loop is utilized to regulate the output voltage V_(O). A second input of the AND gate 91 is coupled to receive the current-control signal S_(I) for achieving output current control. A third input of the AND gate 91 is coupled to receive a maximum-duty signal S_(M). The voltage-control signal S_(V), the current-control signal S_(I) and the maximum-duty signal S_(M) can reset the D flip-flop 95 for shorten the pulse width of the switching signal V_(PWM) no as to regulate the output voltage V_(O) and the output current I_(O). The maximum-duty signal S_(M) is generated by a maximum duty circuit (DMAX) 650. The maximum duty circuit 650 can be utilized to limit the maximum-duty of the switching signal V_(PWM) under 50%.

A positive input of a comparator 700 is coupled to receive a detect signal αV_(IN). A low-voltage threshold V_(TH) is supplied with a negative input of the comparator 700. An enable signal S_(EN) is generated at an output of the comparator 700 by comparing the detect signal αV_(N) with the low-voltage threshold V_(TH). The detect signal αV_(IN) is correlated to the input voltage V_(IN). The output of the comparator 700 generates the enable signal S_(EN) coupled to control an AND gate 710. Two inputs of the AND gate 710 receives the pulse signal PLS and the enable signal S_(EN) respectively. An output of the AND gate 710 generates a sample signal S_(P) coupled to the integrator 500. The detail description for input voltage V_(IN) detection can be found in prior arts “Control method and circuit with indirect input voltage detection by switching current slope detection” U.S. Pat. No. 7,616,461 and “Detection circuit to detect input voltage of transformer and detection method for the same” US 2008/0048633 A1.

The programmable signal V_(CNT) generated at the input terminal VCNT is supplied to a positive input of a buffer amplifier 720. A negative input of the buffer amplifier 720 is connected to its output. A resistor 730 is coupled between the output of the buffer amplifier 720 and a reference voltage device 750. The reference voltage device 750 is connected to the reference signal V_(REF1) to clamp the maximum voltage of the reference signal V_(REF1). The reference voltage device 750 can be implemented by a zener diode. The programmable signal V_(CNT) is coupled to regulate the current-control signal S_(I) of the controller 70 through controlling the reference signal V_(REF1) of a current-loop. Furthermore, the programmable signal V_(CNT) is coupled to control the reference signal V_(REF1) of the current-loop of the controller 70. The switching signal V_(PWM) is modulated in response to the current-control signal S_(I) for regulating the output current I_(O), and the level of the output current I_(O) is correlated to the current-control signal S_(I). In other words, the switching signal V_(PWM) is modulated in response to the reference signal V_(REF1), and the level of the output current I_(O) is correlated to the reference signal V_(REF1).

FIG. 4 is a preferred embodiment of the integrator in accordance with the present invention. An amplifier 510, a resistor 511 and a transistor 512 construct a first V-to-I converter to generate a first current I₅₁₂ in response to the current-waveform signal V_(B). A positive input of the amplifier 510 is supplied with the current-waveform signal V_(B). A negative input of the amplifier 510 is coupled to a source terminal of the transistor 512 and one terminal of the resistor 511. The other terminal of the resistor 511 is coupled to the ground. An output of the amplifier 510 is coupled to a gate terminal of the transistor 512. A drain terminal of the transistor 512 generates the first current I₅₁₂.

Transistors 514, 515 and 519 form a first current mirror for producing a current I₅₁₅ and a current I₅₁₉ by mirroring the first current I₅₁₂. Source terminals of the transistors 514, 515 and 519 of the first current mirror are coupled to the supply voltage V_(CC). Gate terminals of the transistors 514, 515, 519 and drain terminals of the transistors 512, 514 are connected together. Drain terminals of the transistors 515 and 519 generate the current I₅₁₅ and I₅₁₉ respectively. Transistors 516 and 517 form a second current mirror for generating a current I₅₁₇ by mirroring the current I₅₁₅. Source terminals of the transistors 516 and 517 of the second current mirror are coupled to the ground. Gate terminals of the transistors 516, 517 and drain terminals of the transistors 516, 515 are connected together. A drain terminal of the transistor 517 generates the current I₅₁₇.

An amplifier 530, a resistor 531 and a transistor 532 form a second V-to-I converter for generating a second current I₅₃₂ in response to the current-waveform signal V_(A). A positive input of the amplifier 530 is supplied with the current-waveform signal V_(A). A negative input of the amplifier 530 is coupled to a source terminal of the transistor 532 and one terminal of the resistor 531. The other terminal of the resistor 531 is coupled to the ground. An output of the amplifier 530 is coupled to a gate terminal of the transistor 532. A drain terminal of the transistor 532 generates the second current I₅₃₂. Transistors 534 and 535 form a third current mirror for producing a current I₅₃₅ by mirroring the second current I₅₃₂. Source terminals of the transistors 534 and 535 of the third current mirror are coupled to the supply voltage V_(CC). Gate terminals of the transistors 534, 535 and drain terminals of the transistors 532, 534 are connected together. A drain terminal of the transistor 535 generates the current I₅₃₅.

Transistors 536 and 537 develop a fourth current mirror for producing a current I₅₃₇ in response to the current I₅₃₅ and the current I₅₁₇. Source terminals of the transistors 536 and 537 of the fourth current mirror are coupled to the ground. Gate terminals of the transistors 536, 537 and drain terminals of the transistors 536, 535 are connected together. The drain terminal of the transistor 536 and a drain terminal of the transistor 537 generate a current I₅₃₆ and the current I₅₃₇ respectively. The current I₅₃₆ can be expressed by I₅₃₆=I₅₃₅−I₅₁₇. The geometric size of the transistor 536 is twice the size of the transistor 537. Therefore the current I₅₃₇ is the current I₅₃₆ divided by 2. Transistors 538 and 539 form a fifth current mirror for generating a current I₅₃₉ by mirroring the current I₅₃₇. Source terminals of the transistors 538 and 539 of the fifth current mirror are coupled to the supply voltage V_(CC). Gate terminals of the transistors 538, 539 and drain terminals of the transistors 538, 537 are connected together. A drain terminal of the transistor 539 generates the current I₅₃₉. The drains of the transistor 519 and the transistor 539 are coupled together for generating the average current signal I_(AVG) by summing the current I₅₁₉ and the current I₅₃₉. A current feedback signal V_(X) is therefore generated at the drain terminals of the transistor 519 and the transistor 539. The resistor 511, the resistor 531 and a capacitor 570 determine the time constant of the integrator 500, and the resistor 531 is correlated to the resistor 511.

A switch 550 is coupled between the drain terminal of the transistor 519 and the capacitor 570. The switch 550 is controlled by the discharge-time signal S_(DS) and turned on only during the period of the discharge-time of the secondary side switching current I_(S). A transistor 560 is coupled to the capacitor 570 in parallel to discharge the capacitor 570. The transistor 560 is turned on by the clear signal CLR. The integrator 500 further includes a sample-and-hold circuit formed by a sample switch 551 and an output capacitor 571. The sample switch 551 is coupled between the capacitor 570 and the output capacitor 571. The switch 551 controlled by the sample signal S_(P) serves to periodically sample the voltage across the capacitor 570 to the output capacitor 571. The current signal V_(Y) is therefore generated across the output capacitor 571. The sample-and-hold circuit is coupled to sample the current feedback signal V_(X) for generating the current-control signal S_(I) (as shown in FIG. 3). The current feedback signal V_(X) is correlated to the switching current I_(P) of the offline transformer 10 (as shown in FIG. 1). In other words, the sample-and-hold circuit is coupled to sample the current input signal V_(IP) (as shown in FIG. 1) for generating the current-control signal S_(I). As shown in FIG. 3, the sample-and-hold circuit will stop to sample the current feedback signal V_(X) once the input voltage V_(IN) of the drive circuit is lower than the low-voltage threshold V_(TH). In other words, the sample-and-hold circuit will stop to sample the current feedback signal V_(X) once the AC input V_(AC) (as shown in FIG. 2) is lower than the low-voltage threshold V_(TH). The AND gate 710 generates the sample signal S_(P) for sampling of the current feedback signal V_(X).

FIG. 5 shows a preferred embodiment of the maximum duty circuit 650 in accordance with the present invention. The maximum duty circuit 650 includes an inverter 670, a transistor 671, a current source 675, a capacitor 680 and a comparator 690. A gate terminal of the transistor 671 receives the switching signal V_(PWM) through the inverter 670. The switching signal V_(PWM) is coupled to control the transistor 671. The current source 675 is coupled between the supply voltage V_(CC) and a drain terminal of the transistor 671. A source terminal of the transistor 671 is coupled to the ground. The capacitor 680 is connected between the drain terminal of the transistor 671 and the ground. The transistor 671 is coupled to the capacitor 680 in parallel to discharge the capacitor 680 when the switching signal V_(PWM) is disabled. The current source 675 is connected to the supply voltage V_(CC) and is used to charge the capacitor 680 when the switching signal V_(PWM) is enabled. The current source 675 and the capacitance of the capacitor 680 determine the pulse-width and the amplitude of the voltage across the capacitor 680. A negative input of the comparator 690 is coupled to the drain terminal of the transistor 671 and the capacitor 680. A reference signal V_(REF3) is supplied to a positive input of the comparator 690. An output of the comparator 690 generates the maximum duty signal S_(M). To set up the reference signal V_(REF3) appropriately, the maximum duty circuit 650 can be utilized to limit the maximum-duty of the switching signal V_(PWM) under 50%.

FIG. 6 is another preferred embodiment of the controller 70 in accordance with the present invention. The controller 70 generates the switching signal V_(PWM) coupled to switch the offline transformer 10 for generating the current input signal V_(IP) (as shown in FIG. 1). A positive input of a buffer amplifier 780 receives the current input signal V_(IP). A negative input of the buffer amplifier 780 is coupled to its output. A voltage-to-current converter 800 receives the programmable signal V_(CNT) to generate a programmable current I_(CNT). A resistor 790 is coupled between the output of the buffer amplifier 780 and the output of the voltage-to-current converter 800. The resistor 790 and the output of the voltage-to-current converter 800 are further coupled to the input of the waveform detector 300. The programmable current I_(CNT) is further coupled to the current-sense terminal VS (as shown in FIG. 1) via the resistor 790 and the buffer amplifier 780 for modulating the current input signal V_(IP). Hence, the programmable signal V_(CNT) generated at the input terminal VCNT is coupled to modulate the current input signal V_(IP). Referring to the FIG. 3, the current input signal V_(IP) is further coupled to generate the current-control signal S_(I). The current input signal V_(IP) correlated to the switching current I_(P) of the offline transformer 10 and the programmable signal V_(CNT). The switching signal V_(PWM) is controlled in response to the current-control signal S_(I), thus the level of the output current I_(O) is correlated to the current-control signal S_(I).

FIG. 7 shows a preferred embodiment of the voltage-to-current converter 800 in accordance with the present invention. The voltage-to-current converter 800 comprises an amplifier 810, a resistor 825, a transistor 820, a first current mirror formed by transistors 830, 831, a current source 850, a second current mirror formed by transistors 832, 833. A positive input of the amplifier 810 receives the programmable signal V_(CNT). A negative input of the amplifier 810 is coupled to a source terminal of the transistor 820 and one terminal of the resistor 825. The other terminal of the resistor 825 is coupled to the ground. An output of the amplifier 810 is coupled to a gate terminal of the transistor 820. A drain terminal of the transistor 820 is coupled to the first current mirror and generates a current I₈₂₀.

The first current mirror generates a current I₈₃₁ by mirroring the current I₈₂₀. Source terminals of the transistors 830 and 831 of the first current mirror are coupled to the supply voltage V_(CC). Gate terminals of the transistors 830, 831 and drain terminals of the transistors 830, 820 are connected together. A drain terminal of the transistor 831 generates the current I₈₃₁. The second current mirror is coupled to the drain terminal of the transistor 831 to generate a current I₈₃₃ by mirroring the current I₈₃₁. Source terminals of the transistors 832 and 833 of the second current mirror are coupled to the ground. Gate terminals of the transistors 832, 833 and drain terminals of the transistors 832, 831 are connected together. A drain terminal of the transistor 833 generates the current I₈₃₃. The current source 850 is coupled from the supply voltage V_(CC) to the drain terminal of the transistor 833. The drain terminal of the transistor 833 further outputs the programmable current I_(CNT). As shown in FIG. 6, the programmable current I_(CNT) is to modulate the current input signal V_(IP). The current input signal V_(IP) is correlated to the switching current I_(P) of the offline transformer 10 and the programmable signal V_(CNT).

It will be apparent to those skilled in the art that various modifications and variations can be made to the structure of the present invention without departing from the scope or spirit of the invention. In view of the foregoing, it is intended that the present invention cover modifications and variations of this invention provided they fall within the scope of the following claims and their equivalents. 

What is claimed is:
 1. A LED drive circuit comprising: a controller generating a switching signal coupled to switch a primary winding of a magnetic device for generating an output current to drive a plurality of LEDs, in which the controller is a primary-side controller; and a programmable signal coupled to regulate a current-control signal of the controller; wherein the switching signal is modulated in response to the current-control signal for regulating the output current, and the level of the output current is correlated to the current-control signal.
 2. The LED drive circuit as claimed in claim 1, wherein the switching signal is coupled to control a switching current of the magnetic device; the magnetic device is an offline transformer.
 3. The LED drive circuit as claimed in claim 1, wherein the programmable signal is generated in response to an AC input of the LED drive circuit.
 4. The LED drive circuit as claimed in claim 1, wherein the controller comprises a sample-and-hold circuit to sample a current feedback signal for generating the current-control signal; the current feedback signal is correlated to a switching current of the magnetic device; wherein the sample-and-hold circuit stops to sample the current feedback signal once an AC input of the LED drive circuit is lower than a low-voltage threshold.
 5. The LED drive circuit as claimed in claim 1, further comprising a maximum duty circuit for limiting a maximum-duty of the switching signal.
 6. The LED drive circuit as claimed in claim 5, wherein the maximum-duty of the switching signal is limited under 50%.
 7. A LED circuit comprising: a controller generating a switching signal coupled to switch a primary winding of a magnetic device for generating an output current to drive a plurality of LEDs, in which the controller is a primary-side controller; and a programmable signal coupled to control a reference signal of a current-loop of the controller; wherein the switching signal is modulated in response to the reference signal, and the level of the output current is correlated to the reference signal.
 8. The LED circuit as claimed in claim 7, wherein the magnetic device is an offline transformer.
 9. The LED circuit as claimed in claim 7, wherein the programmable signal is generated in response to an AC input of the LED circuit.
 10. The LED circuit as claimed in claim 7, wherein the controller comprises a sample-and-hold circuit to sample a current feedback signal; the current feedback signal is correlated to a switching current of the magnetic device; wherein the sample-and-hold circuit stops to sample the current feedback signal once an input voltage of the LED circuit is lower than a low-voltage threshold.
 11. The LED circuit as claimed in claim 7, further comprising a maximum duty circuit for limiting a maximum-duty of the switching signal.
 12. The LED circuit as claimed in claim 11, wherein the maximum-duty of the switching signal is limited under 50%.
 13. A LED driver comprising: a controller generating a switching signal coupled to switch a transformer for generating a current input signal coupled to the controller and an output current connected to drive a plurality of LEDs; and a programmable signal coupled to modulate the current input signal; wherein the current input signal is further coupled to generate a current-control signal, the current input signal is correlated to a switching current of the transformer; the switching signal is controlled in response to the current-control signal, and the level of the output current is correlated to the current-control signal.
 14. The LED driver as claimed in claim 13, wherein the controller is a primary-side controller that is coupled to switch a primary winding of the transformer.
 15. The LED driver as claimed in claim 13, wherein the controller comprises a sample-and-hold circuit coupled to sample the current input signal; the sample-and-hold circuit stops to sample the current input signal once an input voltage of the LED driver is lower than a low-voltage threshold.
 16. The LED driver circuit as claimed in claim 13, further comprising a maximum duty circuit for limiting a maximum-duty of the switching signal.
 17. The LED driver as claimed in claim 16, wherein the maximum-duty of the switching signal is limited under 50%. 